Circuitry for a low internal voltage

ABSTRACT

A technique provides an on-chip voltage to a core portion of an integrated circuit by way of a conversion transistor. The on-chip voltage may be a reduced internal voltage, less than the VCC of the integrated circuit. In an embodiment, the layout (or physical structure) of the conversion transistor is distributed surrounding the core portion. By providing the core with a reduced voltage, the integrated circuit may interface with other integrated circuits compatible with different voltage levels.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.09/499,166, filed Nov. 24, 1999 now U.S. Pat. No. 6,414,518, which is acontinuation of U.S. patent application Ser. No. 08/863,876, filed May27, 1998, now U.S. Pat. No. 6,025,737, which claims priority to U.S.provisional application No. 60/018,465, filed May 28, 1996; No.60/018,494, filed May 28, 1996; No. 60/018,510, filed May 28, 1996; No.60/022,837, filed Jul. 31, 1996; No. 60/031,617, filed Nov. 27, 1996;and No. 60/046,810, filed May 2, 1997. All references cited above and inthis application are incorporated by reference in their entirety for allpurposes.

BACKGROUND OF THE INVENTION

The present invention relates to the field of integrated circuits, andmore specifically, to improving the interfacing of integrated circuit ina mixed-voltage environment.

The integrated circuit business and semiconductor industry arecontinually driven to reduce cost, reduce power, and improveperformance. The integrated circuit products include microprocessors,memories, programmable logic, programmable controllers, applicationspecific integrated circuits, and many other types of integratedcircuits. Price reduction is strongly driven by migrating products toscaled processes, which reduce die sizes and increase yields. Powerreduction has been achieved by circuit design techniques, powermanagement schemes, and parasitic scaling, among other factors.Performance improvement has resulted from design techniques, processenhancements, and parasitic scaling, among other factors.

Process technology is improving. Resulting from the continual scalingand shrinking of device geometries, device sizes and dimensions requirethe operating voltages to be scaled. Operating voltages have been scaleddown from 5 volts to 3.3 volts. This has resulted in the need formixed-voltage-mode systems. That is, integrated circuits will need tointerface with various operating voltages. And, further reductions areexpected in the future. This industry provides products and printedcircuit boards (PCBs) that utilize both 3.3-volt and 5-volt integratedcircuits and devices. It is expected that there may be a considerabletransition period for the standard power supply to switch from onevoltage level to a lower voltage level.

Process scaling is the dominant method of reducing the die cost. Thecost is achieved by receiving higher yields associated with smaller diesizes. Presently, power supply voltages are being reduced as the scalingprogresses towards device dimensions that necessitate the reduction ofvoltage differences across these dimensions.

All manufacturers have not switched over to the lower power supply,simultaneously. Thus the scaling of the operating voltage has resultedin creating a multiple voltage mode industry. Integrated circuitcompanies must provide products capable of addressing the needs duringthis intermediate phase before the industry transitions to a singlelower power supply voltage. It is expected that this industry willrequire some time to successfully transition over to the lower powersupply.

As can be seen, an improved technique of fabricating, and operatingintegrated circuits is needed to meet these demands. These integratedcircuits should interact with devices that are designed to operate ateither the standard or the new lower power supply. The integratedcircuit should also provide a cost reduction path to customers thatcontinue to design 5-volt-only systems. Integrated circuits shouldprovide the manufacturer with the flexibility to chose the market tosupport with a minimum cost and the shortest time to market.

SUMMARY OF THE INVENTION

The present invention is a technique of interfacing an integratedcircuit in a mixed-voltage mode environment. In particular, theintegrated circuit is fabricated using technology compatible with aninternal supply voltage. Externally, the integrated circuit willinterface with an external supply voltage, above the internal supplyvoltage. The input and output signals to and from the integrated circuitwill be compatible with the external supply level.

An integrated circuit of the present invention will include conversioncircuit for converting a voltage at a level of external supply voltageto a level of the internal supply voltage. In one embodiment, theconversion circuitry uses negative feedback, and is self-regulating.This internal supply voltage will be used to power the internal deviceson the integrated circuit. The integrated circuit will containconversion circuitry to convert output signals to be compatible with theexternal supply voltage. Also, the integrated circuit will also be ableto accept input voltages compatible with the external supply voltage.The integrated circuit will appear to a user and other integratedcircuits as though it were manufactured using technology compatible withthe external supply voltage. The present invention is a useful techniquefor providing backward compatibility of a process technology.

The present invention may be used in an integrated circuit havingseparated noisy and quiet supplies. For example, the I/O drivers may becoupled to a noisy supply while the conversion circuit is coupled to thequiet supply. This will help noise from the I/O drivers from couplinginto the core of the integrated circuit.

A layout of the conversion circuitry is compact and also spreads currentflow and heat distribution evenly around the integrated circuit. Thishelps prevent the formation of localized “hot spot” areas.

More specifically, an integrated circuit of the present inventionincludes an output driver coupled to a first voltage supply; a levelshifter circuit coupled to a second voltage supply; and a voltage downconverter circuit, coupled to the second voltage supply. The voltagedown converter generates a first voltage supply having a voltage levelbelow the second voltage supply. Circuitry in a core of the integratedcircuit is coupled to the first voltage supply.

Other objects, features, and advantages of the present invention willbecome apparent upon consideration of the following detailed descriptionand the accompanying drawings, in which like reference designationsrepresent like features throughout the figures.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a digital system incorporating aprogrammable logic device integrated circuit;

FIG. 2 is a block diagram showing an architecture for a programmablelogic device;

FIG. 3 is a simplified block diagram of a logic array block (LAB) of aprogrammable logic device;

FIG. 4 shows an option for an integrated circuit of the presentinvention providing a single voltage supply solution;

FIG. 5 shows an option for an integrated circuit of the presentinvention providing the capability to tolerate and interface in a mixedvoltage environment;

FIG. 6 shows an option for an integrated circuit of the presentinvention providing the ability to interface with a supply voltage abovethat for a core of the integrated circuit;

FIG. 7 is a flow diagram illustrating a technique of fabricating anintegrated circuit capable of interfacing in a mixed voltageenvironment;

FIG. 8 shows a circuit diagram of an output driver;

FIG. 9A shows a circuit diagram of an output driver, tolerant of highvoltage at an I/O pad, which has dual power supply pins;

FIG. 9B shows a circuit diagram of an alternative circuit embodiment ofa pull-down driver;

FIG. 10A shows a circuit diagram of a high-voltage tolerant outputdriver having a well bias generator;

FIG. 10B shows an implementation of an input buffer;

FIG. 10C shows an implementation of a buffer having a programmable inputthreshold trip point using programmable options;

FIG. 10D shows a buffer with a programmable input threshold configuredto shift the trip point up;

FIG. 10E shows a buffer with a programmable input threshold configuredto shift the trip point down;

FIG. 10F shows another implementation of a buffer having a programmableinput threshold trip point;

FIG. 10G shows an alternative circuit configuration for the buffercircuitry in FIG. 10F;

FIG. 10H shows a further input buffer implementation;

FIG. 10I shows another input buffer implementation including a halflatch;

FIG. 11 shows a circuit diagram of another high-voltage tolerant outputdriver having a well bias generator;

FIG. 12 shows a circuit diagram of a further embodiment of the highvoltage tolerant output driver having a well bias generator;

FIG. 13 shows a circuit diagram of a technique of interfacing anintegrated circuit using a voltage down converter circuit;

FIG. 14 shows an integrated circuit and a layout overview of componentsof the voltage down converter;

FIG. 15 shows a simplified layout diagram of fingers of a device of thevoltage down converter;

FIG. 16 shows a layout of a portion of the voltage down converter,including voltage clamping devices;

FIG. 17 shows a layout of a portion of the voltage down converters,including an inverting amplifier circuit;

FIG. 18 shows a circuit diagram of a voltage down converter and aspecific implementation of an inverting amplifier;

FIG. 19 shows a further embodiment of a voltage down converter where theinverting amplifier may be logically controlled;

FIG. 20A shows a voltage down converter circuit using a biasing currentnetwork;

FIG. 20B shows an alternative embodiment of a voltage down convertercircuit using a biasing current network;

FIG. 21 shows a circuit diagram of a level shifter circuit;

FIG. 22 shows a schematic of a circuitry for interfacing low voltageinternal circuitry with higher voltage external circuitry;

FIG. 23 shows a specific embodiment of a level shifter circuit; and

FIG. 24 shows an embodiment of an isolation device.

DETAILED DESCRIPTION

FIG. 1 shows a block diagram of a digital system within which thepresent invention may be embodied. The system may be provided on asingle board, on multiple boards, or even within multiple enclosures.FIG. 1 illustrates a system 101 in which a programmable logic device 121may be utilized. Programmable logic devices (sometimes referred to as aPALs, PLAs, FPLAs, PLDs, EPLDs, EEPLDs, LCAs, or FPGAs), are well-knownintegrated circuits that provide the advantages of fixed integratedcircuits with the flexibility of custom integrated circuits. Suchdevices allow a user to electrically program standard, off-the-shelflogic elements to meet a user's specific needs. See, for example, U.S.Pat. No. 4,617,479, incorporated herein by reference for all purposes.Such devices are currently represented by, for example, Altera's MAX®series of PLDs and FLEX® series of PLDs. The former are described in,for example, U.S. Pat. Nos. 5,241,224 and 4,871,930, and the Altera DataBook, June 1996, all incorporated herein by reference. The latter aredescribed in, for example, U.S. Pat. Nos. 5,258,668, 5,260,610,5,260,611, and 5,436,575, and the Altera Data Book, June 1996, allincorporated herein by reference for all purposes. Logic devices andtheir operation are well known to those of skill in the art.

In the particular embodiment of FIG. 1, a processing unit 101 is coupledto a memory 105 and an I/O 111 and incorporates a programmable logicdevice (PLD) 121. PLD 121 may be specially coupled to memory 105 throughconnection 131 and to I/O 111 through connection 135. The system may bea programmed digital computer system, digital signal processing system,specialized digital switching network, or other processing system.Moreover, such systems may be designed for a wide variety ofapplications such as, merely by way of example, telecommunicationssystems, automotive systems, control systems, consumer electronics,personal computers, and others.

Processing unit 101 may direct data to an appropriate system componentfor processing or storage, execute a program stored in memory 105 orinput using I/O 111, or other similar function. Processing unit 101 maybe a central processing unit (CPU), microprocessor, floating pointcoprocessor, graphics coprocessor, hardware controller, microcontroller,programmable logic device programmed for use as a controller, or otherprocessing unit. Furthermore, in many embodiments, there is often noneed for a CPU. For example, instead of a CPU, one or more PLDs 121 maycontrol the logical operations of the system. In some embodiments,processing unit 101 may even be a computer system. Memory 105 may be arandom access memory (RAM), read only memory (ROM), fixed or flexibledisk media, PC Card flash disk memory, tape, or any other storageretrieval means, or any combination of these storage retrieval means.PLD 121 may serve many different purposes within the system in FIG. 1.PLD 121 may be a logical building block of processing unit 101,supporting its internal and external operations. PLD 121 is programmedto implement the logical functions necessary to carry on its particularrole in system operation.

FIG. 2 is a simplified block diagram of an overall internal architectureand organization of PLD 121 of FIG. 1. Many details of PLD architecture,organization, and circuit design are not necessary for an understandingof the present invention and such details are not shown in FIG. 2.

FIG. 2 shows a six-by-six two-dimensional array of thirty-six logicarray blocks (LABs) 200. LAB 200 is a physically grouped set of logicalresources that is configured or programmed to perform logical functions.The internal architecture of a LAB will be described in more detailbelow in connection with FIG. 3. PLDs may contain any arbitrary numberof LABs, more or less than shown in PLD 121 of FIG. 2. Generally, in thefuture, as technology advances and improves, programmable logic deviceswith greater numbers of logic array blocks will undoubtedly be created.Furthermore, LABs 200 need not be organized in a square matrix; forexample, the array may be organized in a five-by-seven or atwenty-by-seventy matrix of LABs.

LAB 200 has inputs and outputs (not shown) which may or may not beprogrammably connected to a global interconnect structure, comprising anarray of global horizontal interconnects (GHs) 210 and global verticalinterconnects (GVs) 220. Although shown as single lines in FIG. 2, eachGH 210 and GV 220 line may represent a plurality of signal conductors.The inputs and outputs of LAB 200 are programmably connectable to anadjacent GH 210 and an adjacent GV 220. Utilizing GH 210 and GV 220interconnects, multiple LABs 200 may be connected and combined toimplement larger, more complex logic functions than can be realizedusing a single LAB 200.

In one embodiment, GH 210 and GV 220 conductors may or may not beprogrammably connectable at intersections 225 of these conductors.Moreover, GH 210 and GV 220 conductors may make multiple connections toother GH 210 and GV 220 conductors. Various GH 210 and GV 220 conductorsmay be programmably connected together to create a signal path from aLAB 200 at one location on PLD 121 to another LAB 200 at anotherlocation on PLD 121. A signal may pass through a plurality ofintersections 225. Furthermore, an output signal from one LAB 200 can bedirected into the inputs of one or more LABs 200. Also, using the globalinterconnect, signals from a LAB 200 can be fed back into the same LAB200. In specific embodiments of the present invention, only selected GH210 conductors are programmably connectable to a selection of GV 220conductors. Furthermore, in still further embodiments, GH 210 and GV 220conductors may be specifically used for passing signal in a specificdirection, such as input or output, but not both.

The PLD architecture in FIG. 2 further shows at the peripheries of thechip, input-output drivers 230. Input-output drivers 230 are forinterfacing the PLD to external, off-chip circuitry. FIG. 2 showsthirty-two input-output drivers 230; however, a PLD may contain anynumber of input-output drivers, more or less than the number depicted.Each input-output driver 230 is configurable for use as an input driver,output driver, or bidirectional driver.

FIG. 3 shows a simplified block diagram of LAB 200 of FIG. 2. LAB 200 iscomprised of a varying number of logic elements (LEs) 300, sometimesreferred to as “logic cells,” and a local (or internal) interconnectstructure 310. LAB 200 has eight LEs 300, but LAB 200 may have anynumber of LEs, more or less than eight. In a further embodiment of thepresent invention, LAB 200 has two “banks” of eight LEs for a total ofsixteen LEs, where each bank has separate inputs, outputs, controlsignals, and carry chains.

A general overview of LE 300 is presented here, sufficient to provide abasic understanding of the present invention. LE 300 is the smallestlogical building block of a PLD. Signals external to the LAB, such asfrom GHs 210 and GVs 220, are programmably connected to LE 300 throughlocal interconnect structure 310, although LE 300 may be implemented inmany architectures other than those shown in FIGS. 1-3. In oneembodiment, LE 300 of the present invention incorporates a functiongenerator that is configurable to provide a logical function of a numberof variables, such a four-variable Boolean operation. As well ascombinatorial functions, LE 300 also provides support for sequential andregistered functions using, for example, D flip-flops.

LE 300 provides combinatorial and registered outputs that areconnectable to the GHs 210 and GVs 220, outside LAB 200. Furthermore,the outputs from LE 300 may be internally fed back into localinterconnect structure 310; through local interconnect structure 310, anoutput from one LE 300 may be programmably connected to the inputs ofother LEs 300, without using the global interconnect structure's GHs 210and GVs 220. Local interconnect structure 310 allows short-distanceinterconnection of LEs, without utilizing the limited global resources,GHs 210 and GVs 220. Through local interconnect structure 310 and localfeedback, LEs 300 are programmably connectable to form larger, morecomplex logical functions than can be realized using a single LE 300.Furthermore, because of its reduced size and shorter length, localinterconnect structure 310 has reduced parasitics compared to the globalinterconnection structure. Consequently, local interconnect structure310 generally allows signals to propagate faster than through the globalinterconnect structure.

FIGS. 4-6 illustrate a technique of interfacing integrated circuitsincluding programmable logic devices and field programmable gate arraysto other integrated circuits. With improvements in process technology,integrated circuits use lower power supply voltages such as 3.3 volts or2.5 volts, or even lower. However, integrated circuits made with theseprocesses should remain compatible with previous generation integratedcircuits. For example, a 3.3-volt integrated circuit may need to be usedon a printed circuit board with 5-volt integrated circuits. The 3.3-voltintegrated circuit will need to have the proper supply and inputvoltages for operation. Also, the 3.3-volt integrated circuit shouldsupply or generate the proper output voltages for interfacing with theother integrated circuits. Proper interfacing of the integrated circuitsis essential for proper functional operation. Further, properinterfacing will prevent undesirable conditions, such as overstressingthe devices, avoiding possible high current or latch-up conditions, andother similar concerns. This will improve device longevity.

Using a technique of the present invention, integrated circuits such asprogrammable logic devices and field programmable gate arrays may befabricated with mixed-mode capability. Such an integrated circuit willbe capable of being configured to interface with a variety of integratedcircuits operating at similar and different voltage levels. By selectingand programming the appropriate programmable options, the integratedcircuit will be able to interface with integrated circuits using thesame supply voltage, lower supply voltage, and higher supply voltage.

In a preferred embodiment, the programmable options are implemented byway of metal options selected during processing by selecting and usingthe appropriate masks. For example, a mixed mode integrated circuitmanufactured using 3.3-volt technology may have three options.

In a first option (illustrated by FIG. 4), the integrated circuit iscompatible with 3.3-volt technology. Specifically, input signals fromother integrated circuits should be 3.3-volt compatible. The powersupply will be 3.3 volts. And the output signal will provide 3.3-voltdrive capability.

More specifically, as shown in FIG. 4, a core 405 and an interface 411of the integrated circuit will operate with a 3.3-volt supply. Further,interface 411 will be compatible with input signals from 3.3-voltcircuits, and will generate output signals for interfacing with 3.3-voltcircuits. For a programmable integrated circuit such as a PLD or FPGA,core 405 may include the LABs, LEs, programmable interconnect includingGVs, GHs, and local interconnect. In contrast, interface 411 wouldinclude dedicated input buffers, dedicated output buffers, outputdrivers, input-output buffers, and associated circuitry.

In a second option (illustrated in FIG. 5), the integrated circuit iscapable of tolerating 5-volt input signals. The power supply will be 3.3volts. And the output signal will provide 3.3-volt drive capability. Inthis case, as shown in FIG. 5, the supply voltage for core 405 andinterface 411 will be 3.3 volts. Interface 411 will be tolerant tovoltages from 5-volt integrated circuits. Interface 411 will generate3.3-volt compatible output.

In a third option (illustrated in FIG. 6), the integrated circuit willtolerate 5-volt input signals. The power supply will be 5 volts. And theoutput signal will provide 5-volt compatible drive capability. As anexample, the voltage level for a logic high at the output will be about5 volts—VTN or above. Please note, even in this case, the integratedcircuit will be manufactured using 3.3-volt technology. As shown in FIG.6, the power supply is 5 volts. This voltage is converted using on-chipcircuitry to a lower voltage of 3.3 volts. This conversion may beperformed using a voltage down converter (VDC) 610. The lower voltage issupplied to the circuitry in core 405 and interface 411. Interface 411is capable of tolerating 5-volt input signals. Further, in interface411, the core 3.3-volt signals may be converted to 5-volt output signalsby circuitry such as level-shifting predrivers. The circuits used toperform the conversion in the interface are connected to the 5-voltsupply voltage.

Using the technique of the present invention, there may be more or lessthan three options. An integrated circuit may have, for example, anycombination of two of the modes described above. Further, there may beadditional options, other than those described above. As an example,there may be low power version and high power version of a chip, wherethis is selectable via programmable options. Specific implementations ofsome of the above options are discussed in further detail below.

In the present invention, the circuitry for implementing the threeoptions resides on the integrated circuit. Specifically, in the abovecase of three options, the circuitry necessary for the first option,second option, and third option are on-chip. Then, by appropriatelyconnecting the appropriate circuitry by programmable options (e.g.,programmable links, programmable cells, metal mask options), a mode ordesign is selected for that integrated circuit chip. Moreover, usingthis technique, circuitry on the integrated circuit is shared among themultiple options so that silicon area is conserved. For example, in aprogrammable logic device, a 5-volt tolerant or 3.3-volt specificinput/output (I/O) interface may be programmably selected to beconnected to programmable logic core. This programmable logic core wouldinclude the LABs and LEs and programmable interconnect.

There are many techniques for implementing the programmable optionsfeature of the present invention besides mask programmable options.These include, and are not limited to, laser programmable options,fuses, antifuse, in-system programmable (ISP) options, reprogrammablecells such as EEPROM, Flash, EPROM, and SRAM, and many others.

The voltage levels given above are merely for the purpose of example.The present invention may be easily applied to any mixed voltage levelsituation involving at least two different voltage levels. For example,one of the voltage levels may be 3.3 volts while another voltage levelis 2.5 volts.

Furthermore, the internal circuitry (e.g., core 405) will be compatiblewith a VCCint voltage. There may be separated external supply pins tothe integrated circuit. For example, there may be a noisy supply, VCCNand a quiet supply VCCQ. These will be discussed further below.

For example, for a 2.5-volt technology core, when VCCQ is 2.5 volts andVCCN is 2.5 volts, the first option is selected for an integratedcircuit that will be tolerant to 2.5-volt external signals. When VCCQ is2.5 volts and VCCN is 2.5 volts, the second option is selected for anintegrated circuit that will be tolerant to 3.3-volt external signals,or signals above 3.3 volts. The degree of tolerance to external signalsof 3.3 volts or above may be dependent on numerous factors including theprocess technology used, thickness of the oxide for the devices andtransistors, and many other considerations. When VCCQ is 3.3 volts andVCCN is 3.3 volts or less, the third option is selected, and theintegrated circuit will be tolerant to external signals up to VCCN. WhenVCCQ is 2.5 volts and VCCN is less than 2.5 volts, the first or secondoption may be selected. An integrated circuit with the first option willbe tolerant to external signals up to VCCN. An integrated circuit withthe second option will be tolerant to external signals up to 3.3 voltsor above, depending on process technology considerations such as oxidethickness and others.

Similarly, for a 3.3-volt technology core, when VCCQ is 3.3 volts andVCCN is 3.3 volts, the first option is selected for an integratedcircuit that will be tolerant to 3.3-volt external signals. When VCCQ is3.3 volts and VCCN is 3.3 volts, the second option is selected for anintegrated circuit that will be tolerant to 5-volt external signals.When VCCQ is 5 volts and VCCN is 5 volts or less, the third option isselected, and the integrated circuit will be tolerant to externalsignals up to VCCN. When VCCQ is 3.3 volts and VCCN is less than 3.3volts, the first or second option may be selected. An integrated circuitwith the first option will be tolerant to external signals up to VCCN.An integrated circuit with the second option will be tolerant to theexternal signals to 3.3 volts or above, depending on process technologyconsiderations such as oxide thickness and others.

For a 5-volt technology core, when VCCQ is 5 volts and VCCN is 5 volts,the first option is selected for an integrated circuit that will betolerant to 5-volt external signals. When VCCQ is 5 volts and VCCN isless than 5 volts, the first or second option may be selected. Anintegrated circuit with the first option will be tolerant to externalsignals up to VCCN. An integrated circuit with the second option will betolerant to external signals up to 5 volts.

Integrated circuits may be fabricated according to the technique asshown in FIG. 7. A step 705 provides an integrated circuit corecompatible with an internal supply voltage. For example, this internalsupply voltage may be 3.3 volts. The integrated circuit core may includethe programmable logic (e.g., LABs, LEs, look-up tables, macrocells,product terms) in a PLD or FPGA.

A step 710 provides a first interface option (e.g., see FIG. 4), whichmay provide an interface for the integrated circuit designed to handleinput signals from external circuits compatible with the internalvoltage supply level (e.g., 5 volts) and generate output signals forexternal circuits compatible with the internal supply voltage level. Forexample, using this first interface option, a 3.3-volt only integratedcircuit may be manufactured. This corresponds to an integrated circuitas shown in FIG. 4.

A step 715 provides a second interface (e.g., see FIG. 5), which mayprovide an interface for the integrated circuit designed to handle inputsignals from external circuits compatible with another external supplyvoltage level (e.g., 5 volts) and generate output signals for externalcircuits compatible with the internal supply voltage level (e.g., 3.3volts). For example, using the second interface option, a 3.3-voltintegrated circuit that is tolerant of 5-volt input signals may bemanufactured. This corresponds to an integrated circuit as shown in FIG.5

A step 720 provides a third interface (e.g., see FIG. 6), which mayprovide an interface for the integrated circuit designed to handle inputsignals from external circuits compatible with the external supplyvoltage level (e.g., 5 volts) and generate output signals for externalcircuits compatible with the external supply voltage level (e.g., 5volts). For example, using the third interface option, a 5-volt externalintegrated circuit may be manufactured using 3.3-volt process and devicetechnology. The internal circuitry will operate at 3.3 volts. Thiscorresponds to the integrated circuit as shown in FIG. 6. The secondsupply voltage may be generated on-chip.

In a preferred embodiment, the circuitry to implement these threeinterface options are formed on the same integrated circuit orsemiconductor body as the core.

A step 725 involves selectively coupling the first interface, secondinterface, or third interface to the core. Step 725 may be performed byselectively programming the integrated circuit, such as by metalmasking, e-beam lithography, programming laser fuses, programmablefuses, antifuse, electrically erasable programmable cells, and manyothers. The selected interface option will be programmablyinterconnected with the core. Circuitry to implement the options may beresident on the integrated circuit; however, the circuitry is not neededto perform a particular interface option will be disabled. Furthermore,the same circuitry may be “reused” in multiple interface options. Thiswill aid in providing an even more compact layout.

Using the technique of the present invention, an integrated circuit maybe easily manufactured to be compatible with different operatingenvironments, without specifically designing an individual integratedcircuit for each specific case. This leads to reduced research anddevelopment and production costs. This also reduces the risk of holdingan excess inventory in unneeded integrated circuit types. In particular,integrated circuits where an interface option has not yet been selected,may be selectively fabricated or programmed with the appropriateinterface option as needed. This allows a much faster response time forfabricating the desired integrated circuits to meet rapidly changingmarket conditions.

FIG. 8 shows an output driver which may be used in interface 411 of theintegrated circuit. Such an output driver may be used in one of theinterface options of the integrated circuit. In particular, thiscircuitry may be used in the implementation of the first option shown inFIG. 4. The output driver includes a pull-up driver 810 and a pull-downdriver 815. In this embodiment, pull-up driver 810 is a PMOS transistorand pull-down driver 815 is an NMOS transistor. Pull-up driver 810 iscoupled between a supply 817 and a pin (or pad) 820. Pin 820 maysometimes be referred as an I/O pad as it may be used for input oroutput, or both. Pull-down driver 815 is coupled between pin 820 and asupply 822. Supply 817 is typically VDD or VCC and supply 822 istypically VSS.

In operation, the output driver will generate a logic high, logic low,or be tristated (i.e., high impedance state) depending on the logicsignals at PU and PD. PU is coupled to a gate of pull-up driver 810 andPD is coupled to a gate of pull-down driver 815. When PU is a low and PDis a low, the pin will be driven high (to the level of VCC). When PU ishigh and PD is high, the pin will be driven low (to the level of VSS).When PU is high and PD is low, the pin will be tristated. Pin 820 istypically coupled to an input buffer (not shown) for the inputting oflogical signals into the integrated circuit and the core. Pin 820 may beused as an input when the output buffer is placed in tristate, or mayalso be used to feed back signals from the output buffer into theintegrated circuit.

However, the output driver circuit shown in FIG. 8 is not tolerant tohigh voltages, and would not be useful in the case where input voltagesare from an integrated circuit having a supply voltage above a level offirst supply 817. For example, when the output buffer is tristated,signals are input to the input buffer (not shown) via pin 820. If firstsupply 817 is 3.3 volts, then when interfacing a 5-volt integratedcircuit, pin 820 may potentially be 5 volts or above. A 5-volt inputwould represent a logic high input. This voltage may even go above 5volts during transitions due to glitches and switching noise. This posespotential problems.

An I1 current sneak path (or leakage path) will occur when the VPIN (thevoltage level at the pin) goes above 3.3 volts+|VTP|. VTP is thethreshold voltage of pull-up driver 810. Furthermore, in an embodiment,pull-up driver 810 is a PMOS transistor and formed in an n-well on ap-type substrate. In that case, there is a parasitic diode 830 betweenpin 820 and first supply 817. Parasitic diode 830 represents the diodebetween the p-diffusion used to form the drain and the n-well region,which is connected to first supply 817. Therefore, an I2 current sneakpath will also occur when the VPIN goes above 3.3 volts+Vdiode. Vdiodeis the turn-on or forward voltage (VF) of the diode.

Sneak current paths I1 and I2 will allow the first supply (VCC) to rise.If VCC rises greater than absolute maximum allowable levels and remainat those levels for a longer than acceptable time, the device will haveoxide reliability issues. Therefore, it is undesirable for the outputbuffer shown in FIG. 8 to interface with voltage levels above firstsupply 817.

FIG. 9A shows an output driver circuit which is tolerant to high-voltageinputs at pin 820. This circuitry may be used for the second interfaceoption described above and shown in FIG. 5. In FIG. 9A, the outputdriver has a separate 5-volt supply pin 910 and 3.3-volt supply pin 817.When a separate supply pin is not available or desirable, this 5-voltsupply voltage may be internally generated by a voltage pump or othersimilar means. For example, an internal voltage of 5 volts may begenerated from the 3.3-volt supply. The n-well for pull-up driver 810 (aPMOS transistor) is connected to node 910. This n-well (which will be at5 volts) will prevent the I2 current path discussed above.

Furthermore, circuitry is coupled to PU to prevent the I1 current path.The circuitry used to bias node PU includes PMOS transistors MP1 and MP2and NMOS transistor MN1. MP1 is coupled between supply 910 and PU andMN1 is between PU and VSS. MP2 is between supply 910 and a gate of MP1(node 915). Body connections for MP1 and MP2 are coupled to supply 910.A chain of inverters X00, X01, and X02 feed through an NMOS passgatetransistor 920 into the gate of MP1. A gate of MN1 is coupled to anoutput of inverter X00. An input node In inputs inverter X00.

When an input node IN is low, PU will be low since MN1 is on and MP1 isoff. In this case, the sneak current paths are not a concern sincepull-up driver 810 will be on. In the case when IN is high, PU will beat the level of supply 910 (e.g., 5 volts). When PU is 5 volts, I1 willnot conduct unless VPIN is 5 volts+|VTP|. Therefore, there will be no I1path when pin 820 is at 5 volts. PU is 5 volts because node 915 is lowand supply 910 (5 volts) is passed through MP1 to PU. The n-wells of MP1and MP2 are connected to supply 910 in order to prevent problems such aslatch-up and to minimize the body effect. The n-well of MP3 may also becoupled to supply 910 for this purpose.

Pass transistor 920 also serves to isolate inverter X02 from node 915.Even when the node 915 is above the voltage of supply 817 (e.g., 3.3volts), pass transistor 920 (MN2) will limit the voltage at an output ofinverter X02 to the voltage level of supply voltage 817 less a VTN(i.e., threshold voltage of pass transistor 920). This will preventunduly overstressing the devices used to form inverter X02.

In a preferred embodiment, MN3, MN1, MP1, MP2, and MP3 are thick oxidedevices. MN2 may also be a thick oxide device, which would ensurereliability under the condition when supply 817 is off and nodes 910 and915 are at 5 volts. Thick oxide devices are transistors which havethicker gate oxide than the thin gate oxide used for other transistors.For example, a thin oxide device may have an oxide thickness of about 70Angstroms. A thick oxide device can typically tolerate greater voltagestress than a thin oxide device. For example, a thick oxide device maybe able to handle 5-volt or greater stress. A typical thick oxidethickness may be about 140 Angstroms. By using thick oxide devices, thiswill reduce oxide stress for these devices when interfacing withvoltages above supply 817 at pin 820. Also, these thick oxide deviceswill be less likely to breakdown due to high voltages at pin 820.Therefore, the overall longevity and operation of the integrated circuitis enhanced.

FIG. 9B shows an alternative pull-down circuitry for an output buffersuch as shown in FIG. 9A. Transistors 942 and 944 would be used insubstitution for MN3. A gate of transistor 942 is coupled to firstsupply 817. A gate of transistor 944 is coupled to PD. Transistors 942and 944 are NMOS devices, and are thin oxide devices.

Although this pull-down circuitry is formed using thin oxide devices, itwill be tolerant to high voltages at pin 820. Specifically, highvoltages will be divided between the two transistors so neither deviceis subject to too high a voltage which would damage the devices.Transistor 942 limits a voltage at node 946 to VDD−VTN. The circuitry inFIG. 9B may be useful in cases to provide tolerance to high voltages,but where it is undesirable to use a thick oxide device, or when thickoxide devices are not available.

However, there may be some disadvantages when using two thin oxidedevices compared to one thick oxide device. For example, more siliconarea may be used by having two devices instead of one. Also, theperformance when using two devices may be slightly less due to increasedparasitics and other similar considerations.

For the above discussion, supply voltage 910 was described as 5 voltswhile supply voltage 817 was described as 3.3 volts. These values weregiven only for the purpose of example. As would be apparent to thoseskilled in the art, the circuitry would operate and function analogouslyfor different, specific voltages where supply voltage 910 is abovesupply voltage 817. For example, supply voltage 910 may be 3.3 volts andsupply voltage 817 may be 2.5 volts.

FIG. 10A shows another output driver (or output buffer) which will allowinterfacing with high voltages at pin 820. In this embodiment, a wellbias generator 1002 is used to bias an n-well and a gate of pull-updriver 810. The output driver circuitry in FIG. 10A is similar to thoseshown in FIGS. 8 and 9. FIG. 10A also shows an input buffer XINV3 whichis coupled to pin 820 for coupling signals to the core of the integratedcircuit. A further discussion of the input buffer is presented below andin conjunction with FIGS. 10B, 10C, and 10D.

The embodiment in FIG. 10A has a supply voltage 817. The integratedcircuit may have a “noisy” power supply (i.e., VCCN) and a “quiet” powersupply (i.e., VCCQ). Both the noisy and quiet supplies may be connectedto the same voltage level. However, the noisy power supply would beconnected to a separate pin from a quiet power supply. On the integratedcircuit, the noisy power supply would be connected to circuitry whichgenerates or is subject to noise, while the quiet power supply would beconnected to relatively quiet circuitry. By separating the powersupplies in this fashion, the circuitry connected to the quiet powersupply will be isolated somewhat from switching and other types of noisepresent on the noisy power supply.

The noisy supply would be connected to relatively noisy circuitry suchas output drivers (e.g., supply voltage 817 may be a noisy supply). Forexample, output drivers generate noise from ground bounce. Further, inan integrated circuit such as shown in FIG. 5, the circuitry ininterface 411 would generally be connected to the noisy supply sincethese circuits are typically considered “noisy.” The circuits in core405 would be connected to the quiet supply since these circuits aretypically considered “quiet.” This will tend to help prevent noise fromcoupling into the core of the integrated circuit.

In certain embodiments, such as will be described below, it may bedesirable to couple certain devices (e.g., transistor 920 or others) tothe second supply voltage, which would be a quiet supply voltage. Then,supply voltage 817 would be the noisy supply. In this embodiment, thedevices are coupled to the same supply voltage 817, which may be a noisyor quiet supply. In a specific embodiment, supply voltage 817 will be anoisy supply voltage while the core of the integrated circuit is coupledto a quiet supply voltage.

Well bias generator 1002 includes transistors M7 and M8 which arecoupled between supply 817 and a bias output node 1010. A gate oftransistor M7 is coupled to a node 1015. A gate of transistor M8 iscoupled to bias output node 1010.

Transistor M9 and M10 are coupled between a node 1015 and the biasoutput node 1010. A gate of transistor M9 is coupled to supply 817. Agate of transistor M10 is coupled to bias output node 1010.

Node 1015 is connected through a resistor R3 to pin 820. Resistor R3 maybe used to provide electrostatic discharge (ESD) protection for devicesM7, M9, M10, and XINV3 from pin 820. However, resistor R3 may beoptionally omitted depending on the particular embodiment. Othertechniques for ESD protection may be used.

Additionally, a transistor M11 is coupled between the bias output node1010 and PU. An inverter chain including inverters XINV1 and XINV2 drivethrough pass transistor 920 to PU. Pass transistor 920 may besubstituted with other pass gate structures. Pass transistor 920 may besubstituted with a transmission gate, CMOS transmission gate (includingan NMOS transistor and a PMOS transistor), two of more transistors inseries, and many other specific circuit implementations. An output 1020of XINV1 drives a gate of transistor M11.

In a preferred embodiment transistors M7, M8, M9, M10, and M11 are PMOStransistors. N-well connections for transistors M7, M8, M9, M10, and M11are coupled to bias output node 1010.

In operation, well-bias generator 1002 generates a bias output voltage1010 which is used to prevent currents I1 and I2 shown in FIG. 8. Asshown in FIG. 10A, bias output node 1010 is coupled to the n-well ofpull-up driver 810. Furthermore, bias output node 1010 can be coupled ordecoupled to the gate of pull-up driver 810 depending on the conditions.

More specifically, when PU is low, the output of inverter XINV1 will behigh. In this case, transistor M11 will be off and effectively decoupledfrom node PU. This is the case when pin 820 is driven to a logic high.The I1 and I2 current paths are not of a concern.

On the other hand, when PU is high, the output of inverter XINV1 will below. In this case, transistor M11 will be on. Transistor M11 willeffectively couple bias output node 1010 to the gate of PU. Essentially,the gate of PU will track the voltage at gate bias output node 1010 inorder to prevent current path I1 described above.

Voltage bias generator 1002 will be described in connection with thevoltage conditions at pin 820. In particular, bias output node 1010 willbe VCC (i.e., the level at supply voltage 817) when pin 820 is in arange from ground to about VCC−|VTP|. VCC is the voltage at supply 817,and |VTP| is the threshold voltage for a PMOS transistor. Bias outputnode 1010 is coupled to VCC through transistor M7, which will be in aconducting or on state. Under these conditions, voltage bias generator1002 prevents I1 and I2 current paths. Specifically, the gate and n-wellof pull-up driver 810 will be biased to VCC. Since VPIN (i.e., thevoltage level at pin 820) will be less than VCC, I1 and I2 will be zero.

In the case when pin 820 is above about VCC−|VTP| but below about VCC,transistor M7 will be off. Bias output node 1010 will be held to aboutVCC−|VTP| through transistor M8. Note that transistor M8 may besubstituted with a diode or similar device or component. For example,the p-n junctions of the transistors 810, M7, and M8 form such a diode.This would serve a similar function of maintaining bias output node 1010at around VCC−VF. VF is the forward voltage of the diode. Under theseconditions, voltage bias generator 1002 keeps the gate and n-well ofpull-up driver 810 biased properly. The I1 and I2 current paths are notof concern.

In the case when pin 820 goes above VCC, but below about VCC+|VTP|, biasoutput node 1010 will be about VPIN−|VTP|, where VPIN is a voltage levelat pin 820. Bias output node 1010 will be held at this level throughtransistor M10. Transistor M10 acts like a diode, analogous totransistor M8. Similarly, transistor M8 may also be substituted with adiode structure or other device or component as discussed in the casefor transistor M8. For example, such a diode is present in the p-njunctions of transistors M9 and M10. Under these conditions, the gateand n-well of pull-up driver 810 will be about VPIN−|VTP|. The I1 and I2current paths will not be of concern. If |VTP| were slightly greaterthan the VF of diode 830 (see FIG. 8), then there may be a relativelysmall current I2. However, I2 would be zero when |VTP| is less than theVF of diode 830.

In the case when pin 820 goes above about VCC+|VTP|, bias output node1010 will be VPIN. VPIN will be passed through transistor M9. TransistorM9 will be in a conducting state under these conditions. Under theseconditions, the gate and n-well of pull-up driver 810 will be the sameas VPIN. In this case, current paths I1 and I2 will also not occur.

Therefore, as described above, voltage bias generator 1002 prevents theI1 and I2 sneak current paths in the case when a voltage above supplyvoltage 817 is placed at pin 820. For example, an integrated circuitwith a 3.3-volt supply voltage may be driven with a 5-volt inputvoltage. The output driver circuitry shown in FIG. 10A may be used inimplementing the second option (shown in FIG. 5) for a mixed voltagemode capable integrated circuit.

In a preferred embodiment for the circuitry in FIG. 10A, as discussedpreviously, pull-down driver 815 and transistor M11 should be thickoxide devices. This is to ensure the gate oxide reliability at differentvoltage stress conditions. For pull-down driver 815, a stress conditionoccurs when pin 820 is at about 5 volts and node PD is grounded. Fortransistor M11, a stress condition occurs when pin 820 is at about 5volts, which makes node 1010 about 5 volts, and node 1020 will be atabout ground. Further, the |VTP| for thick oxide devices may differ fromthe |VTP| for thin oxide devices. Therefore, transistors M7 and M9 mayalso be thick oxide devices in order to ensure they have a similar |VTP|as pull-up driver 810. This is important in order that voltage biasgenerator 1002 track the characteristics of pull-up driver 810 properly.However, if |VTP| for thin oxide devices is less than that for thickoxide devices, then transistors M7 and M9 may be thin oxide devices.This is because the difference between the voltage at pin 820 and PUwill be less |VTP| of pull-up transistor 810. This ensures no I1 currentpath.

Transistors M3, M8, and M10 may also be thick oxide devices. Dependingon the process technology used, there may be advantages to using thickoxide devices such as providing improved gate oxide stress tolerance,tracking of device parameters between devices, and other factors.

Further, devices M3, M7, M8, M9, and pull-up driver 810 may also bethick oxide devices to improve their oxide reliability. For example, anoxide stressing condition may occur when supply 817 is off, and pin 820,node 1015, and node PU are at 5 volts.

In an embodiment of the present invention, the control electrode of passtransistor 920 may be coupled to a noisy supply (VCCN) or the quietsupply (VCCQ). This connection may be made using a programmable option,such as by programmable link, fuse, programmable bit, and metal mask,just to name a few of the possible techniques. When the controlelectrode of transistor 920 is coupled to VCCQ, the other devices arecoupled to supply 817, which would be VCCN.

In a situation when a voltage level of the VCCN is below that of VCCQ(e.g., VCCN is less than 3.3 volts and VCCQ is 3.3 volts; VCCN is lessthan 2.5 volts and VCCQ is 2.5 volts), then the control electrode oftransistor 920 should be coupled to VCCN. This prevents leakage fromVCCQ to VCCN through transistor 920 since a node 1030 at the output ofXINV2 will be limited to VCCN−VTN; regardless of the voltage level atPU.

Another example where the control electrode of pass transistor 920should be coupled to VCCN is when VCCQ is about 3.3 volts and VCCN isabout 2.5 volts. Under those circumstances, allow for ten percenttolerances on the VCCN and VCCQ, VCCQ may be about 3.6 volts and VCCNmay be about 2.25 volts. If the control electrode of pass transistor 920were coupled VCCQ of 3.6 volts, bias output node 1010 may be 2.25 volts,and node PU will be about VCCQ−VTN, which is approximately 2.6 volts.Then, when node 1020 is zero volts, there will be current flow throughM11, since this device is on. This current flow will be from PU (at 2.6volts) to bias output node 1010 (at 2.25 volts) and to the pin 820. Tominimize this current, M11 may be made into a weak transistor (e.g., bysizing the device).

Another solution is to connect the control electrode of pass transistor920 to VCCN. This will limit the voltage at PU so that nothing will bedriving PU when PU reaches VCCN−VTN.

FIGS. 10B-10I show various implementations of an input buffer of thepresent invention. Input buffer XINV3 in FIG. 10A may be implementedusing these circuit implementations.

FIG. 10B shows an implementation of an input buffer using an invertercircuit configuration. A transistor 1050 and a transistor 1055 arecoupled in series between a positive supply and ground. Controlelectrodes of the two transistors are coupled together, and coupled topin 820. An output 1058 from the inverter is taken from a node betweenthe two inverters. Output 1058 is coupled to drive the core of theintegrated circuit. The positive supply may be VCCQ or VCCint.

In a specific embodiment, transistor 1050 is a p-channel device andtransistor 1055 is an n-channel device. As discussed above, pin 820 maybe subject to voltages above VCCQ or VCCint. For example, pin 820 may beat about 5 volts and VCCQ or VCCint will be about 3.3 volts. In order tominimize oxide stress and improve reliability of the input buffer,transistors 1050 and 1055 may be thick oxide devices, individually or incombination.

Furthermore, the input threshold trip point of the inverter may beprogrammable. The input threshold trip point depends on the ratio of therelative strengths of the ratio of the pull-up transistor 1050 to thepull-down transistor 1055.

Therefore, the trip point may be varied by adjusting the W/L ratio oftransistor 1050 to transistor 1055. The sizes of transistors 1050 and1055 may be adjusted by programmable option. For example, by metal maskoption, the trip point may be adjusted as desired for the intendedapplication.

A programmable threshold input buffer is especially useful for anintegrated circuit which will interface with various voltage suppliesand voltage levels. For example, the input threshold may be adjusted toadapt the integrated circuit for use in 2.5-volt or 1.8-volt supplyenvironments. Furthermore, in situations when VCCQ is above VCCN (e.g.,VCCQ is 3.3 volts and VCCN is 2.5 volts), the input level specificationfor the case when VCC is 2.5 volts may be violated since VCCQ isactually 3.3 volts. A programmable threshold input buffer will be ableto handle the situation to set the input threshold appropriately.

FIG. 10C shows an example of an implementation of the input buffer inFIG. 10B using programmable metal options, such as by selecting anappropriate metal mask. The effective size (or strength) of transistor1050 may be adjusted using transistors 1060 and 1062. Similarly, theeffective size (or strength) of transistor 1055 may be adjusted usingtransistors 1064 and 1066. Transistors 1060, 1062, 1064, and 1066 areprovided in the layout, and are optionally connected in parallel withtransistors 1050 and 1055. Although only a particular number of “option”transistors 1060, 1062, 1064, and 1066 are shown, there may be as manyoption transistors as desired. The option transistors may be of varyingsizes, which may be used to fine-tune the input threshold trip point.

For example, FIG. 10D shows how the input threshold trip point may beshifted up by coupling option transistors 1060 and 1062 in parallel withtransistors 1050. FIG. 10E shows how the input threshold trip point maybe shifted down by coupling option transistors 1064 and 1066 in parallelwith transistor 1055.

In order to provide greater oxide stress tolerance, option transistors1060, 1062, 1064, and 1066 may be thick oxide devices, as was discussedfor transistors 1050 and 1055.

FIG. 10F shows another embodiment of a programmable input thresholdbuffer. This buffer may also be used to implement buffer XINV3 in FIG.10A. The circuitry includes transistors 1050 and 1055, as was discussedfor FIG. 10B. The threshold may be adjusted using transistors 1068 and1070, and transistors 1072 and 1074. There may be additional branches oftransistors, similar to transistors 1068 and 1070, and 1072 and 1074, inparallel with transistors 1050 and 1055. These additional branches oftransistors would allow greater flexibility and precision in theadjustment of the input threshold, similar to the multiple metal optiontransistors in FIG. 10C.

Transistors 1068, 1070, 1072,and 1074 are coupled in series between apositive supply and ground. The positive supply may be VCCQ or VCCint ina multiple positive supply system. A control electrode (or gate) oftransistor 1068 is coupled to a first programmable element PGM1. Controlelectrodes of transistors 1070 and 1072 are coupled to an input pin. Acontrol electrode of transistor 1074 is coupled to a second programmableelement PGM2. The programmable elements may be programmed to represent alogic high or logic low.

The programmable elements may be implemented using mask options. SRAMcells, RAM cells, EPROM cells, EEPROM cells, Flash cells, fusees,antifuses, ferroelectric memory, ferromagnetic memory, and many othertechnologies. For example, PGM1 or PGM2, or both, may be controlled bylogic signals from within a programmable logic device.

By appropriately programming PGM1 and PGM2, the input threshold isshifted up or down. For example, when PGM1 and PGM2 are logic low, theinput threshold trip point is shifted up. When PGM1 and PGM2 are logichigh, the input threshold trip point is shifted down. When PGM1 is logichigh and PGM2 is logic low, the input threshold trip point will not beadjusted. When PGM1 is logic low and PGM2 is logic high, the inputthreshold trip point may be adjusted, depending on the ratio oftransistors 1068 and 1070 to transistors 1072 and 1074. The transistors1070 and 1072 may be thick oxide devices.

FIG. 10G shows an alternative configuration for transistors 1068, 1070,1072, and 1074 of FIG. 10F. The arrangement of the transistors isdifferent but the functionality is similar.

FIG. 10H shows a further embodiment of an input buffer of the presentinverters. The circuitry in FIG. 10H is similar to that in FIG. 10B,with the addition of a transistor 1075 coupled between pin 820 and anode igb at an input of the inverter formed by transistors 1050 and1055. A control electrode of transistor 1075 is coupled to the positivesupply, which may be VCCQ or VCCint in a multiple supply integratedcircuit.

In this embodiment, transistors 1050 and 1055 may be thin oxide deviceswhile transistor 1075 is a thick oxide device. Thick oxide transistor1075 will serve as isolation for thin oxide transistors 1050 and 1055,to minimize stress on the gate oxide of transistors 1050 and 1055.

The buffer circuit in FIG. 10H may be slower than that in FIG. 10B dueto the isolating pass transistor 1075. Furthermore, there may be DCpower consumption. Node igb will be one VTN below VCCQ (i.e., thevoltage at the control electrode of transistor 1075), and there may becurrent flowing through transistors 1050 and 1055 since transistor 1050may still be conducting with VCCQ−VTN at its control electrode.

Furthermore, the input threshold of the buffer may be programmable suchas by using techniques as described above and shown in FIGS. 10F and10G.

FIG. 10I shows another embodiment of an input buffer of the presentinvention. This embodiment shares similarities to the circuits in FIGS.10B and 10H. The circuitry in FIG. 10I further includes a transistor1078 coupled between the positive supply (i.e., VCC, VCCint, or VCCQ)and a node igc. A control electrode of transistor 1078 is coupled to anoutput of the buffer. As in FIG. 10H, transistor 1075 is a thick oxidedevice which isolates thin oxide devices 1050 and 1055 from high voltageoxide stress, as discussed.

Transistors 1078 acts as a p-channel half latch to restore the voltagelevel at node igc to VCCQ when the input (i.e., pin 820) is a logichigh. In FIG. 10I, the control for the half-latch is taken from anoutput of the buffer, however, there are many circuit configurationswhich would accomplish a similar logical function. By ensuring node igcis restarted to VCCQ, this minimizes static or DC power consumptionbecause transistor 1050 will be fully off (as compared to the circuitconfiguration in FIG. 10H). However, transistor 1078 may contribute someDC leakage current at the I/O pin.

Furthermore, the input threshold of the buffer may be programmable suchas by using similar techniques as described above and shown in FIGS. 10Fand 10G.

FIG. 11 is a diagram of a further embodiment of the voltage biasgenerator of the present invention. In FIG. 11, a voltage bias generator1102 is similar to voltage bias generator 1002 of FIG. 10A. Only thedifferences between voltage bias generator 1102 and voltage biasgenerator 1002 will be discussed.

Voltage bias generator 1102 has a bias output node 1110 which is coupledto the n-well of pull-up driver 810. Transistors M7, M8, M9, and M10 areconfigured and operate similarly as the similarly labeled transistors involtage bias generator 1002. These transistors generate the voltage atbias output node 1110.

A voltage at PU is generated by transistors M17, M19, and M11A, incontrast to a single transistor M11 in the embodiment of FIG. 10A.Transistor M17 is coupled between first supply 817 and a node 1120. Acontrol electrode of transistor M17 is coupled to node 1120. TransistorM19 is coupled between node 1015 and node 1120. A control electrode oftransistor M19 is coupled to node 1120. Transistor M11A is coupledbetween node 1120 and PU. A control electrode of transistor M11A iscoupled to an output of inverter XINV1.

In a preferred embodiment, transistors M17, M19, and M11A are PMOSdevices. N-well connections for these transistors are coupled to biasoutput node 1110.

In operation, when PU is low, the output of inverter XINV1 will be high.In this case, transistor M11A will be off and effectively decoupled fromnode PU. This is the case when pin 820 is driven to a logic high. The I1and I2 current paths are not of a concern.

On the other hand, when PU is high, the output of inverter XINV1 will below. In this case, transistor M11A will be on. Transistor M11 willeffectively couple a voltage at node 1120 to node PU. This voltage at PUis used to prevent current path I1 described above. Transistors M17 andM19 will operate analogously to transistor M8 and M10 to bias thevoltage at PU. The operation of this circuitry will be described inrelation to the voltage at pin 820.

When VPIN is less than about VCC, the circuitry will drive PU to aboutVCC−|VTP| through transistors M17 and M11A. This is analogous to theoperation of transistor M8, which was described above. Therefore, the I1current path will be prevented under these conditions.

When VPIN is above about VCC, the circuitry will drive PU to aboutVPIN−|VTP|. This is analogous to the operation of transistor M10, whichwas described above. In this case, the I1 current path will also beprevented since PU will be within a |VTP| of VPIN.

Therefore, voltage bias generator 1102 in FIG. 11 operates similarly tovoltage bias generator 1002 in FIG. 10A. This is because transistor M17serves a similar function as transistor M8, and transistor M19 serves asimilar function as transistor M10. For voltage bias generator 1002, asimilar voltage that is provided at node 1120 is taken from bias outputnode 1010 instead. The circuit configuration in FIG. 10A is preferredsince fewer transistors are required. Otherwise, the operation of bothvoltage bias generator circuits is largely functionally equivalent.

In different embodiments of the present invention, some of the devicesmay be thick oxide devices as was discussed in FIG. 10A. For example, aswas discussed above, pull-down driver 815 and transistor M11 should bethick oxide devices to improve their oxide stress reliability. To ensurea similar VTP, transistors M7 and M9 may be thick oxide devices.Transistors M8, M3, M17, and pull-up driver 810 may also be thick oxidedevices in order to improve their oxide reliability. M10 and M19 may bethick oxide devices.

FIG. 12 shows another embodiment of a voltage bias generator 1202 of thepresent invention. This voltage bias generator shares similarities withthose shown in FIGS. 10 and 11. The differences between the circuitswill be described below.

Voltage bias generator 1202 is similar to voltage bias generator 1002 ofFIG. 10A. Transistors M7, M8, M9, and M10 are configured and operatesimilarly to the similarly labeled transistors in FIG. 10A. A biasoutput node 1210 is coupled to the n-well connection of pull-up driver810. Voltage bias generator 1202 will prevent the I2 current path, aswas previously described.

In this embodiment, a transistor M14 is coupled between pin 820 and PU.A control electrode of transistor M14 is coupled to first supply 817. Apass transistor 1227 is coupled in parallel with pass transistor 920. Acontrol electrode of pass transistor 1227 is coupled to pin 820. In apreferred embodiment, transistor M14 and pass transistor 1227 are PMOStransistors. N-well connections for transistor M14 and pass transistor1227 are coupled to bias output node 1210.

In operation, when VPIN is less than about VCC+|VTP|, transistor M14will not conduct, and decouples pin 820 from PU. Also, when VPIN is lessthan about VCC−|VTP|, transistor 1227 will be on and allow a full-raillogic high voltage (e.g., 3.3 volts when VCC is 3.3 volts) to pass toPU. These transistors ensure the I1 current path will not be of aconcern. These transistors ensure the voltage level at PU will be withinabout a |VTP| of VPIN, and consequently, there will be no I1 currentpath.

When VPIN goes above VCC−|VTP|, PU will track VPIN through transistorM14. Transistor M14 and pass transistor 1227 will not conduct. Morespecifically, the voltage at VPIN will be about VPIN−|VTP|. Under theseconditions, the I1 current path will not be a concern since the VPINwill be within about a |VTP| of the voltage at PU.

Further, in an alternative embodiment of the present invention, passtransistor 920 is a native device. A native device is a transistor whichhas no or minimal VT adjust implant so that the transistor's thresholdvoltage (VTnative) is about zero volts or slightly above. For example,VTnative may be about 0.2 volts. In the case when VTnative is less than|VTP|, pass transistor 1227 may be omitted from the circuitry, thussaving some silicon area.

The circuitry would still function properly because the voltage at PUwill be at least about VCC−VTnative. Specifically, when VCC is driventhrough pass transistor 920, the voltage at PU will be aboutVCC−VTnative. This ensures VPIN will be within a |VTP| of the voltage atPU. Therefore, current path I1 will be prevented.

In different embodiments of the present invention, some of the devicesmay be thick oxide devices as was discussed for FIGS. 10A and 11. Forexample, as was discussed above, pull-down driver 815 and transistor1227 should be thick oxide devices to improve their oxide stressreliability. To ensure a similar VTP, transistors M7 and M9 may be thickoxide devices. Transistors M8, M3, M10, M14, and pull-up driver 810 mayalso be thick oxide devices in order to improve their oxide reliability.

FIG. 13 is a block diagram of an embodiment of the option (such as shownin FIG. 6) where the external power supplies provided to the integratedcircuit will be at a higher voltage than the supply voltage used by theinternal circuitry. Further, the interface circuits will interface withthe high voltage level. For example, the internal circuitry may operatewith a 3.3-volt supply (VCCint) while the external supply voltage(VCCext) is 5 volts. The input and output signals to the chip will be5-volt compatible signals.

As shown in FIG. 13, the integrated circuit has a core 1310 which iscoupled to a level shifter (LS) 1317. The core 1310, as discussedearlier, contains the internal circuitry of the integrated circuit whichis not contained in interface 411 (see FIG. 6). For example, in a PLD orFPGA, core 1310 would include LABs, LEs, GVs, GHs, and other componentsand circuitry. In a microprocessor, core 1310 would include registers,adders, ALUs, instruction execution units, and other components.Interface 411 would contain, for example, the circuitry to generateoutput signals for the integrated circuit.

In the embodiment of the present invention shown in FIG. 13, there areseparated quiet and noisy supplies, which were described previously. Aquiet external supply voltage 1335 (i.e., VCCext) is provided to theintegrated circuit. Using a voltage down converter (VDC) 1330, VCCext isconverted to a lower supply voltage 1340 for the circuitry in core 1310.A noisy external supply voltage 1338 (i.e., VCCN) is coupled to an I/Odriver 1323. VCCN may be at the same voltage level as VCCext. VCCN mayalso be at a different voltage level than VCCext. VCCN is used forinterfacing “noisy” circuitry so that noise is not coupled into VCCext.

Furthermore, in the embodiment of FIG. 13, there is also a quiet andnoisy ground supply, VSSQ 1341 and VSSN 1345, respectively. VSSQ iscoupled to core 1310 while VSSN is coupled to I/O driver 1323. A quietground is separated from the noisy ground in order to prevent couplingof noise into the quiet ground.

In other embodiments of the present inventions, there may a single powersupply VCC, separated noisy and quiet power supplies VCCext (or VCCQ)and VCCN, single ground VSS, separated noisy and quiet grounds VSSN andVSSQ, and combinations of these. For example, there may be a singlepower supply VCC coupled to both core 1310 and I/O driver 1323; however,there may be a noisy ground and a quiet ground. There may also be morethan two separate supplies. For example, there may be separated groundsfor different groupings of I/O drivers 1323 across the integratedcircuit.

The number of power supplies used is somewhat dependent on the number ofpins available for the integrated circuit. The number of power supplypins and ground pins available for the integrated circuit depend on thechip's die size, package used, and other considerations.

Level shifter 1317 converts signals from core 1310 into compatiblesignals for an I/O driver 1323. Level shifter 1317 is coupled to VCCext.For example, level shifter 1317 may convert the 3.3-volt logic signal toan equivalent 5-volt logic signal, which is used to drive I/O driver1323. I/O driver 1323 generates 5-volt-compatible logic signals at a pinor pad.

I/O driver 1323 includes an output driver having a pull-up driver and apull-down driver. For example, I/O driver 1323 may include pull-updriver 810 and pull-down driver 815 as shown in FIG. 8.

There is a voltage down converter (VDC) 1330 which converts VCCext intoa VCCint voltage 1340, which is used by the internal circuitry of theintegrated circuit. VCCint is a voltage less than VCCext. VCCint iscoupled to and supplies the supply voltage for the circuitry in core1310 of the chip. Voltage down converter 1330 is on-chip.

For example, VCCext may be 5 volts, the voltage down converter 1330converts this voltage to a VCCint of about 3.3 volts, or possibly evenlower. To users interfacing this integrated circuit, the chip would beappear to be a 5-volt compatible chip, while the internal circuitryoperates at 3.3 volts. Moreover, in a PLD integrated circuit, forexample, core 1310 may have 3.3-volt logic signals which are passedacross a global interconnect through one or more LABs to level shifter1317. Level shifter 1317 converts these logic into 5-volt compatiblesignals that are passed to the outside world.

In the present invention, because of the on-chip voltage down converter,a separate voltage regulator or voltage converter is not necessary. Thissaves space on a printed circuit board.

In voltage down converter 1330, a transistor 1355 is coupled betweenVCCext and VCCint. VCCint is coupled to an inverting amplifier 1360,which is coupled to a control electrode node 1365 of transistor 1355.Electrode node 1365 is clamped to VCCext using two diode-connectedtransistors 1367 and 1369. Depending on the process technology used,transistors 1367, 1369, and 1355 may be thick oxide devices to providefor greater gate oxide reliability. Transistor 1355 may be a thick oxidedevice in order to improve oxide reliability under conditions when node1365 is about 4 volts or above.

Transistors 1367 and 1369 may be substituted with diodes and othersimilar voltage clamping devices. Transistors 1367 and 1369 operate tomaintain electrode node 1365 within about two VTNs of VCCext. Thisminimizes gate oxide stress on transistor 1355. Therefore, in apreferred embodiment, when VCCext is 5 volts, the voltage level atelectrode 1365 should be about 3.4 volts. A voltage level of about 3.4volts, which is relatively close to the VCCint voltage, is desirablesince it allows faster response time for the inverting amplifier toadjust for fluctuations in the voltages. Further, in a specificembodiment, when VCCint is about 3.4 volts, the current throughtransistor 1355 is designed to conduct a relatively small amount ofcurrent. For example, this current may be less than about one milliamp.Depending on the technology (e.g., voltage drop per voltage clamp) usedand the design criteria, there may be more or fewer than two voltageclamps. For example, there may be only one voltage clamp or there may bethree or more voltage clamps.

During operation, the voltage level at VCCint may fluctuate for a numberof reasons including noise, fluctuations at VCCext, and voltage sag whencore 1310 draws a large amount of current, just to name a few. Voltagedown converter 1330 is self-regulating to obtain a relatively stableVCCint. As VCCint drops, inverting amplifier 1360 turns on electrodenode 1365 more strongly, which increases conduction through transistor1355. This increases VCCint. When VCCint is too high, the oppositeeffect occurs. Conduction through transistor 1355 is restricted toreduce VCCint. Therefore, voltage down converter 1330 generates aself-regulating VCCint, regulated using negative feedback.

As discussed above, in one embodiment, VCCint would be around 3.3 volts.And, the circuitry is implemented so that VCCint would not drop below 3volts at predetermined voltage sag conditions. These conditions takeinto consideration the performance of the integrated circuit under theworst case operating conditions and voltages. The performance of theintegrated circuit will also meet or exceed the specifications under theworst case operating conditions. Specifically, under these conditions,the response time for VCCint to the sag conditions does not cause speedor performance degradation since 3 volts would have been one of theworst case operating conditions. (This would be a worst case operatingvoltage.) This will also ensure the circuitry on the integrated circuitwill operate and function properly.

In a preferred embodiment, transistor 1355 is an NMOS transistor.Transistor 1355 is shown as a single device, but may be multiple devicescoupled in parallel. Transistor 1355 should be a rather large device inorder to supply the power requirements of the integrated circuit.

An example of the power requirements is that 2.5 amps may be dynamicallyrequired during operation (i.e., AC switching). A width of transistor1355 may be about 4500 microns. Transistor 1355 may be formed usingabout 750 smaller devices in parallel. Each individual device may be 6microns in width.

In a preferred embodiment, the channel length should be greater thanminimum in order to permit the transistor to handle greater voltagestresses. As a specific example, if the minimum drawn channel length forthe process is 0.6 microns, the drawn channel length for transistor 1355would be about 0.75 microns. This would improve the reliability of thedevice and avoid the effects of electromigration and hot electrondegradation.

In order to distribute this power evenly across the entire integratedcircuit, these individual devices may be evenly distributed surroundingthe core of the integrated circuit as shown in FIG. 14. Transistor gates1425 represent each of the individual gate widths for transistor 1355.These individual gate widths may be referred to as “fingers” oftransistor 1355. Transistor 1355 is fed by VCCext using bus 1430, whiletransistor 1355 supplies VCCint internal to bus 1435.

FIG. 15 shows a more detailed diagram of a layout of an individualtransistor 1510 used to form transistor 1355. Metal-3 buses are used todistribute VCCint and VCCext. Diffusion regions 1515 and 1517 arecoupled to VCCext using metal-1 which is coupled to metal-2 and then tometal-3. Similarly, a diffusion region 1520 is coupled to VCCint.Polysilicon is used to form control electrode (i.e., a gate) 1365 oftransistor 1355. Inverting amplifier 1360 will be coupled to thepolysilicon.

Forming the transistor 1355 as shown in FIGS. 14 and 15 provides certainbenefits including evenly distributing current and power throughout theintegrated circuit. The IR (voltage) drop and turn-on resistance areminimized. This means there will be less chance for generating “hotspots” on the integrated circuit, where a portion of the integratedcircuit is subject to abnormally high temperature compared to the restof the integrated circuit. This is undesirable since the reliability ofthe integrated circuit may be reduced. Also, since the device is formedusing metal fingers, the structure will act analogously to a big heatfin (e.g., heat sink), which draws heat away from the integratedcircuit.

FIG. 16 shows a layout of a portion of transistor 1355. The specificconnections between the geometries and layers are similar to those shownand described for FIG. 15. A plurality of transistor fingers 1610 isused to form this portion of transistor 1355. VCCint is coupled to oneside of transistor 1355. VCCext is coupled to transistor 1355.Furthermore, transistors 1367 and 1369, used for voltage clamping, areshown coupling to electrode 1365.

The portion of the transistor 1355 shown in FIG. 16 may be repeated asmany times as necessary, or as space permits. Note that transistors 1367and 1369 may also be repeated for each grouping of transistor fingers.In this case, there would be multiple instances of transistors 1367 and1369 throughout the integrated circuit. Since each occurrence oftransistors 1367 and 1369 would be distributed around the integratedcircuit, this improves the response time for these devices as theparasitics delays will be less.

FIG. 17 shows another layout of a portion of transistor 1355. FIG. 17shows similar features as FIG. 16. However, inverting amplifier 1360 isshown coupled to electrode node 1365. This structure may be repeatedmany times in an integrated circuit to achieve the desired size fortransistor 1355. Analogous to the discussion for FIG. 16, there may bemultiple instances of inverting amplifier 1360 (coupled in parallel)distributed around the integrated circuit. This also improves theresponse time for the inverting amplifier since the parasitic delays arereduced.

FIG. 18 is a schematic of an implementation of voltage down converter1330. Inverting amplifier 1360 is formed using a first transistor 1805and a second transistor 1810, which are coupled in series between VCCext1335 and VSSQ 1341. An output of inverting amplifier 1360 is taken frombetween first transistor 1805 and second transistor 1810, and coupled tocontrol electrode node 1365. A control electrode of first transistor1805 is coupled to VCCint. Similarly, a control electrode of secondtransistor 1810 is coupled to VCCint.

In a preferred embodiment, first transistor 1805 is a PMOS transistorwhile transistor 1810 is an NMOS transistor. A layout of this embodimentof inverting amplifier 1360 is shown in FIG. 17 (pointed to by referencenumber 1360). Note in this implementation, clamping transistors 1367 and1369 are not shown; however, these devices may be optionally includedfor the reasons discussed above.

FIG. 19 is a schematic of a further embodiment of the present invention.In this embodiment, there are a plurality of inverting amplifiers 1360A,1360B, and 1360C. This schematic represents an implementation whereindividual amplifiers are distributed around the integrated circuit.Inverting amplifiers 1360A, 1360B, and 1360C use similar circuitry.Furthermore, inverting amplifiers 1360A, 1360B, and 1360C are controlledby signals at nodes 1930A, 1930B, and 1930C, respectively.

In this embodiment, an inverting amplifier 1360 (e.g., 1360C) hastransistors 1920, 1922, 1924, and 1926 coupled in series between VCCextand VSSQ. An output of inverting amplifier 1360C is taken from betweentransistors 1922 and 1924, and coupled to control electrode node 1365.Control electrodes of transistor 1922 and 1924 are coupled to VCCint.

A control electrode of transistor 1926 is coupled to a first controlsignal at node 1930C. A control electrode of transistor 1920 is coupledto a second control signal 1935, which is a complement of the firstcontrol signal at node 1930C, generated by buffer 1910C. Specifically,in FIG. 19, buffer 1910C is a CMOS inverter which uses VCCext and VSSQas its supplies.

In operation, inverting amplifier 1360C is turned on or off depending onfirst control signal at node 1930C and second control signal 1935. Whenfirst control signal at node 1930C is a logic high, second controlsignal 1935 is a logic low, inverting amplifier 1360C is enabled andoperates similarly to inverting amplifier 1360 shown in FIG. 18. On theother hand, when first control signal 1930C is a logic low, secondcontrol signal 1935 is a logic high, inverter amplifier 1360C isdisabled and effectively decoupled from electrode node 1365.

Inverting amplifiers 1360A and 1360B operate similarly as described forinverting amplifier 1360C. First control signal at node 1930C may beuseful for controlling the amount of power dissipation in the integratedcircuit since inverting buffers 1360A, 1360B, and 1360C may beselectively turned off.

FIG. 20A shows a diagram of a further embodiment of a voltage downconverter 1330 of the present invention. In this embodiment, transistor1355 is coupled between VCCext 1335 and a first terminal of biasingcurrent network 2001. In this embodiment, transistor 1355 is a PMOStransistor. A control electrode node 1365 of transistor 1355 is coupledto the first terminal of biasing current network 2001. The firstterminal of biasing current network 2001 is coupled to and used forVCCint. Biasing current network 2001 contains circuitry to maintain aconstant current through transistor 1355, which generates a stablevoltage at VCCint. VCCint is coupled to the core of the integratedcircuit at node 1340. Biasing current network 2001 ensures VCCint doesnot charge up to the voltage of VCCext 1335. There are manyimplementations for biasing current network 2001. For example, biasingcurrent network 2001 may implemented using current mirrors, voltageregulators, operational amplifiers, or combinations of these, just toname a few.

FIG. 20B shows a diagram of another embodiment of the present invention.This embodiment is similar to the embodiments shown in FIGS. 13 and 20A.Transistor 1355 in this embodiment is a native device having a thresholdvoltage that is generally below that for an enhancement device. Similarto the embodiment in FIG. 13, a control electrode of transistor 1355 iscoupled to an output of inverter 1360. Inverter 1360 is coupled toVCCint 1365. This embodiment also includes a biasing current network2001. Biasing current network 2001 ensures VCCint does not charge up tothe voltage of VCCext 1335, similar to the embodiment in FIG. 20A.

FIG. 21 shows a schematic of level shifting circuit 1317. A transistor2105 and a transistor 2108 are coupled in series between VCCext andVSSQ. A control electrode of transistor 2105 is coupled to a node 2112.A control electrode of transistor 2108 is coupled to a node 2115, whichis coupled to core 1310. A transistor 2117 is coupled between VCCext andnode 2112. A control electrode of transistor 2117 is coupled to anoutput of level shifting circuit 1317 at a node 2120, which is coupledto I/O driver 1323. A transistor 2115 is coupled between nodes 2115 and2112. A control electrode of transistor 2125 is coupled to VCCint. In analternative embodiment, this voltage may be VCCext if the core is ableto operate with and tolerate a voltage of about VCCext−VTN. In apreferred embodiment, transistors 2105 and 2117 are PMOS transistors,and transistors 2108 and 2125 are NMOS transistors.

In operation, a logic low input at node 2115 will result in a logic highoutput at node 2120. The voltage level for this logic high output atnode 2120 will be VCCext, passed through transistor 2105. Transistors2108 and 2117 will be off. Moreover, for transistor 2117, VCCext atoutput node 2120 feeds back to turn transistor 2117 off completely.

A logic high input at node 2115 will result in a logic low output atnode 2120. Specifically, the voltage level for this logic low output atnode 2120 will be VSSQ, passed through transistor 2108. VSSQ from node2120 will turn transistor 2117 fully on so that node 2112 will be atVCCext. VCCext at node 2112 will turn transistor 2105 fully off. Also,VCCext is isolated from node 2115 since the maximum voltage that can bepassed through transistor 2125 to node 2115 is about VCCint−VTN.

The circuitry in FIG. 21 is an example of a specific implementation of alevel shifting circuit. Other circuit embodiments may also be used.

In a specific embodiment, transistors 2105, 2108, 2117, and 2125 may bethick oxide devices (individually and in combination with another) inorder to ensure oxide reliability. One situation where the oxide isstressed is when VCCext and VCCint are powered up at different times. Inthat case, VCCint may be about ground while VCCext is about 5 volts.When node 2120 is about ground, node 2112 will be at about 5 volts.

FIG. 22 shows a diagram of a circuit implementation for interfacing lowvoltage internal circuitry with higher voltage external circuitry. Thiscircuitry may be used within an option (such as shown in FIGS. 6 and 13)where the integrated circuit provides a high-level output voltage higherthan a supply voltage for the internal circuitry. This circuitry mayalso be used in other options.

The circuitry may be used as an input/output buffer for an integratedcircuit. The circuitry is coupled to a VCC1 supply voltage and a VCC2supply voltage. VCC1 is at a voltage level above VCC2. For example, VCC1may be about 3.3 volts while VCC2 may be about 2.5 volts. VCC2 iscoupled to the internal circuitry of the integrated circuit. VCC2 may beinternally generated, such as by using an on-chip voltage down converteras shown in FIG. 13, or may be provided through an external pin. TheVCC2 voltage may be provided externally from an off-chip voltageregulator or converter, or other voltage generating means (e.g., powersupply, transformer, and others). VCC1 is at the voltage level forexternal interfacing. For example, when VCC1 is 3.3 volts, theintegrated circuit will be able to generate external voltages of about3.3 volts.

The circuitry includes a pull-up driver 2205 coupled in series with apull-down driver 2210, between VCC1 and VSS. A node between pull-updriver 2205 and pull-down driver 2210 is coupled to a pad 2215 forinterfacing to external circuitry. Pad 2215 may also be coupled to aninput buffer 2220 for inputting signals into the integrated circuit.Signals from the output drivers may also be fed back through inputbuffer 2220 into the chip. In a preferred embodiment, pull-up device2205 is a PMOS transistor, which has a body electrode coupled to VCC1.In a case where a voltage level at a pad 2115 may exceed VCC1, afloating well may be needed for pull-up device 2205. A specific floatingwell implementation was previously discussed (e.g., see FIG. 10).Pull-down device 2210 is a NMOS transistor.

The output driver circuitry in FIG. 22 operates analogously to theoutput driver circuitry shown in FIG. 8. A control electrode of pull-updriver 2205 is coupled to a PU signal. The PU signal is generated from asignal from internal circuitry, such as buffer 2223, which is coupled toVCC2. This signal output from buffer 2223 is coupled through levelshifter 2225 to PU. Level shifter 2225 is coupled to VCC1 and performsthe same function as the level shifter 1317 of FIG. 13. Specifically,level shifter 2225 will shift the voltage output level from buffer 2223to one which is compatible with the VCC1 supply voltage.

A buffer 2230 is coupled to PD which is coupled to the control electrodeof pull-down device 2210. Buffer 2230 is coupled to the VCC2 supplyvoltage.

By appropriately controlling the voltages at PU and PD, the outputcircuitry will produce logical output at pad 2215 in the voltage rangebetween VCC1 and VSS. The output at pad 2215 may also be tristate.

To turn on pull-up device 2205, level shifter 2225 will couple VSS toPU. To turn off pull-up device 2205, level shifter 2225 will couple VCC1to pull-up device 2205. When VCC1 is coupled to the control electrode ofpull-up device 2205, there will not be sneak current or leakage path forsimilar reasons as discussed above in connection with FIG. 9A.

Pull-up driver 2205 and the devices used to implement input buffer 2220may be thick oxide devices in order to ensure oxide reliability, asdiscussed above for the implementation shown in FIG. 21. One situation,among others, where this may be necessary is to address the situationwhen VCC1 (external supply) and VCC2 (internal supply) are powered up atdifferent times as discussed above.

There are many techniques for implementing a level shifter. For example,a particular embodiment is shown in FIG. 21. FIG. 23 shows anotherimplementation of a level shifter. In a preferred embodiment, this levelshifter is on the same integrated circuit as the core of the integratedcircuit. This will make more economical use of the printed circuit boardarea. However, an off-chip level shifter may also be used in particularembodiments. For example, by using an on-chip level shifter, aparticular integrated circuit may interface with both low voltage andhigh voltage integrated circuits at the same time.

The circuit configuration in FIG. 23 is a cross-coupled latch 2310 andan isolation device 2315. In one embodiment, isolation device 2315 is anNMOS transistor 2320 having a control electrode coupled to VCC2. A firstterminal of isolation device 2315 is an input 2321 for the levelshifter.

Cross-coupled switch 2310 has a first buffer 2322 which includes a firstpull-up device 2325 and a first pull-down device 2330, coupled in seriesbetween VCC1 and VSS. An input of first buffer 2322 is coupled to asecond terminal 2331 of isolation device 2315. An output 2333 of firstbuffer 2322 is also an output of the level shifter. This output willtypically produce output in the range from VSS to VCC1.

A second buffer 2335 for cross-coupled switch 2310 includes a secondpull-up device 2340 and a second pull-down device 2345, coupled inseries between VCC1 and VSS. An output of second buffer 2335 is coupledto an input of first buffer 2322. Similarly, output 2333 is coupled toan input of second buffer 2335.

In a preferred embodiment, the pull-up devices 2325 and 2335 are PMOSdevices while the pull-down devices 2340 and 2345 are NMOS devices. ThePMOS devices may have a floating well as similarly as described for thePMOS devices in FIG. 10A. Alternatively, the PMOS devices may have asubstrate or well connection to VCC1.

The operation of the level shifter in FIG. 23 is similar to thatdescribed for the circuitry in FIG. 21. Input 2321 will be in a rangefrom VSS to about VCC2. When input 2321 is a low, first buffer 2322 willoutput a logic high which will be at about VCC1 at output 2333. Secondbuffer 2335 will output a logic low of about zero volts. A controlelectrode of second pull-up device 2335 will be at VCC1, which will turnthat device off completely.

When input 2321 is a logic high (e.g., about VCC1), first buffer 2322will output a logic low will be at about VSS at output 2333. Secondbuffer 2335 will output a logic high of about VCC1. Consequently, VCC1will be coupled to a control electrode of first pull-up device 2325,which will turn that device off completely.

Isolation device 2315 prevents a voltage above VCC2−VTN from passing tonode 2321. This prevents high voltages from possibly damaging corecircuitry coupled at node 2321.

FIG. 24 shows a isolation device 2415 which may be substituted forisolation device 2315. An NMOS device 2420 is a thick oxide device witha VTthick. VTthick may be about 1 volt or above. A control electrode ofNMOS device 2420 is coupled to VCC1. With this isolation circuitry, thevoltage at node 2321 will be at most VCC1−VTthick, which should still berelatively safe for interfacing with the low-voltage core circuitry.Further, since device 2420 is a thick device, it will be able totolerate the VCC1 voltage at its control electrode. Whether isolationdevice 2315 or 2415 is used depends on many factors including theprovision of the various devices by the process technology.

The foregoing description of preferred embodiments of the invention hasbeen presented for the purposes of illustration and description. It isnot intended to be exhaustive or to limit the invention to the preciseform described, and many modifications and variations are possible inlight of the teaching above. The embodiments were chosen and describedin order to best explain the principles of the invention and itspractical applications to thereby enable others skilled in the art tobest utilize the invention in various embodiments and with variousmodifications as are suited to the particular use contemplated. It isintended that the scope of the invention be defined by the claimsappended hereto.

What is claimed is:
 1. A circuit comprising: a first PMOS transistorcoupled between a first supply voltage and an output node; a second NMOStransistor coupled between the output node and a ground voltage; and aplurality of threshold adjust circuits, each comprising: a third andfourth PMOS transistor coupled between the first supply voltage and theoutput node; and a fifth and sixth NMOS transistor coupled between theoutput node and ground, wherein gates of the first, second, fourth, andfifth transistors are coupled to an input node, and a gate of thirdtransistor is coupled to a first programmable cell and a gate of thefifth transistor is coupled to a second programmable cell.
 2. Thecircuit of claim 1 wherein by selectively programming the first orsecond programmable cell, an input threshold of the circuit is altered.3. A programmable logic device comprising the circuit of claim
 1. 4. Thecircuit of claim 1 wherein by programming the first and secondprogrammable cells to be logic lows, an input threshold of the circuitis shifted higher.
 5. The circuit of claim 1 wherein by programming thefirst and second programmable cells to be logic highs, an inputthreshold of the circuit is shifted lower.
 6. A programmable logicintegrated circuit comprising: a logic array block, programmablyconfigurable to perform logical functions; an output driver coupled to afirst voltage supply; a voltage down converter circuit, coupled to thesecond voltage supply to generate a third voltage supply having avoltage level below the second voltage supply, wherein the logic arrayblock is coupled to the third voltage supply and the voltage downconverter comprises: a conversion transistor coupled between the secondvoltage supply and the third voltage supply; an amplifier coupled toreceive the third voltage supply and feed back a control signal to theconversion transistor; and an input buffer circuit with a programmablyadjustable input threshold, programmably coupled to the logic arrayblock.
 7. The programmable logic integrated circuit of claim 6 whereinthe input buffer comprises: a first PMOS transistor coupled between afirst supply voltage and an output node; a second NMOS transistorcoupled between the output node and a ground voltage; and a plurality ofthreshold adjust circuits, each comprising: a third and fourth PMOStransistor coupled between the first supply voltage and the output node;and a fifth and sixth NMOS transistor coupled between the output nodeand ground, wherein gates of the first, second, fourth, and fifthtransistors are coupled to an input node, and a gate of third transistoris coupled to a first programmable cell and a gate of the fifthtransistor is coupled to a second programmable cell.
 8. The programmablelogic integrated circuit of claim 7 wherein by programming the first andsecond programmable cells to be logic lows, an input threshold of thecircuit is shifted higher.